Frequency-shifted feedback cavity used as a phased array antenna controller and carrier interference multiple access spread-spectrum transmitter

ABSTRACT

An optical processor for controlling a phased antenna array uses a frequency-shifted feedback cavity (FSFC), which includes a traveling-wave cavity. The FSFC incrementally delays and incrementally frequency shifts optical signals circulating in the traveling-wave cavity. Optical signals coupled out of the FSFC are separated by frequency, hence by delay, and processed to control either or both transmit and receive beam-forming operations. The FSFC provides a receiver with multiple receive signals which have incremental values of frequency. Each frequency corresponds to an incremental time sampling of optical signals input into the FSFC. Transmit signals coupled out of the FSFC have frequency and phase relationships that result in short time-domain pulses when combined. Controlling modulation and frequency of the transmit signals achieves carrier interference multiple access, a new type of spread-spectrum communications.

RELATED APPLICATIONS

This application is a Divisional of U.S. patent application Ser. No.09/393,431, filed Sep. 10, 1999, now U.S. Pat. No. 6,888,887, which is aDivisional of U.S. patent Ser. No. 09/022,950, filed Feb. 12, 1998, nowU.S. Pat. No. 5,955,992, each of which is incorporated herein byreference.

BACKGROUND OF THE INVENTION

I. Field of the Invention

The present invention relates to wireless communication and radarsystems. More specifically, the present invention relates to a novel andimproved antenna array processor that controls beam-forming and scanningoperations and that also introduces a new spread spectrum technique.

II. Description of the Related Art

Multiple access communication techniques include time-division multipleaccess (TDMA), frequency division multiple access (FDMA), amplitudemodulation, and spread spectrum. Spread spectrum techniques provide someimprovements over the other multiple access techniques depending on thetype of spread spectrum used. Spread spectrum techniques are based onthe principle of expanding a transmitted baseband signal in frequency.This achieves superior interference-rejection by utilizing high processgain to reduce noise and interference in the received signal.

There are four basic types of spread spectrum. Frequency-hopping spreadspectrum (FHSS) is a well-known technique that provides effectiverejection of narrow-band jamming interference and mitigates near-farinterference. Chirped FM spread spectrum is a technique used primarilyin radar systems. Orthogonal frequency division multiplexing (OFDM) isused to spread high data-rate information streams into multiple lowdata-rate streams carried on separate carrier frequencies. Directsequence CDMA (DS-CDMA) is particularly useful in multiple accesscommunication systems because it allows for very efficient use of thefrequency spectrum and provides for improved frequency reuse. There arealso hybrid techniques that combine various aspects of the four basicspread spectrum types. Most notable are frequency-hopped directsequence, time-division direct sequence, and orthogonal frequency CDMA(also known as multifrequency CDMA or MF-CDMA).

Frequency reuse is the process of using the same frequency in twoseparate geographic regions for two distinct communication links.Frequencies can be reused provided that the two regions are attenuatedor isolated from each other by a minimum value for signal rejection byuser receivers in each region. U.S. Pat. No. 4,901,307 describes theprocess of creating marginal isolation, which provides an increase infrequency reuse in DS-CDMA systems. In DS-CDMA, even small reductions inthe overall power level of the system allow for increased systemcapacity. One particularly effective method for creating isolation andimproving frequency reuse is spatial division multiple access (SDMA).SDMA applications to multiple access communication systems includingadaptive array processing are discussed in U.S. Pat. No. 5,642,353, U.S.Pat. No. 5,592,490, U.S. Pat. No. 5,515,378, and U.S. Pat. No.5,471,647. In addition to frequency reuse, antenna arrays also provideincreased processing gain and improved interference rejection.

The advantage to using adaptive antenna arrays for DS-CDMAcommunications is that adaptive antenna arrays could provide significantimprovements in range extension, interference reduction, and capacityincrease. To identify a particular user, a DS-CDMA system demodulatesWalsh codes after converting the received signal from RF to digital.Therefore, an adaptive antenna array requires information about the usercodes from CDMA radio, or it needs to demodulate many different incomingRF signals to track mobile users. These methods are complex processesand are more difficult to implement than the tracking of users innon-CDMA systems. Major changes in CDMA radio architecture are requiredto implement adaptive array processing. These changes may be the majorobstacle for adaptive array deployment in the near future.

Phased array antenna systems employ a plurality of individual antennasor subarrays of antennas that are separately excited to cumulativelyproduce an electromagnetic wave that is highly directional. The radiatedenergy from each of the individual antenna elements or subarrays is of adifferent phase so that an equiphase beam front, or the cumulative wavefront of electromagnetic energy radiated from all of the antennaelements in the array, travels in a selected direction. The differencein phase or timing between the antenna's activating signals determinesthe direction in which the cumulative wave front from all of theindividual antenna elements is transmitted. Analysis of the phases ofreturn beams of electromagnetic energy detected by the individualantennas in the array similarly allows determination of the directionfrom which a return beam arrives.

Beamforming, which is the adjustment of the relative phase of theactuating signals for the individual antennas, can be accomplished byelectronically shifting the phases of the actuating signals. Beamformingcan also be performed by introducing a time delay in the differentactuating signals to sequentially excite the antenna elements whichgenerate the desired direction of beam transmission from the antenna.However, phase-based electronically controlled phased array systems arerelatively large, heavy, complex, and expensive. These electronicsystems require a large number of microwave components (such as phaseshifters, power splitters, and waveguides) to form the antenna controlsystem. This arrangement results in a system that is relatively lossy,electromagnetically sensitive, hardware-intensive, and has a narrowtunable bandwidth.

Optical control systems can be advantageously used to create selectedtime delays in actuating signals for phased array systems. Suchoptically generated time delays are not frequency dependent and thus canbe readily applied to broadband phased array antenna systems. Forexample, optical signals can be processed to establish the selected timedelays between individual signals, thus causing the desired sequentialactuation of the transmitting antenna elements. The optical signals canthen be converted to electrical signals, such as by a photodiode array.Different types of optical architectures have been proposed to processoptical signals that generate selected delays. Examples of thesearchitectures are fiber optic segments of different lengths for routingthe optical signals; deformable mirrors for physically changing thedistance light travels along a reflected path before being converted toan electrical signal; and free space propagation based delay lines,which typically incorporates polarizing beam splitters and prisms.

U.S. Pat. No. 5,117,239 and U.S. Pat. No. 5,187,487 describe a systemthat creates a cluster of optical beams coupled intoindividually-controlled pixels of a spatial light modulator (SLM). TheSLM provides selectable phase shifts to each of the beams. Some opticaldelay devices, such as U.S. Pat. No. 5,461,687, utilize the refractiveproperties of different wavelengths of light to provide individuallycontrolled phase shifting of wavelength-multiplexed light. Althoughoptical processing offers great improvements over radio frequency (RF)and digital array processing, current optical processing approachesmerely replace microwave components with optical components withoutreducing the complexity of the system. For example, an optical systemhaving a number N of array elements requires N phase-shifters and Nassociated phase-shifter control systems. Some devices, such as Rotmanlenses, are designed to reduce or eliminate the need for adjustablephase shifters. However they increase system complexity and size byintroducing complex elements and systems as well as by introducingadditional detectors.

Several optical systems that exhibit unusual properties have been built,but their application to phased array signal processing had beenoverlooked. In the Optics Letters article “Broadband Continuous WaveLaser,” applicant described a laser design that utilizes atraveling-wave frequency-shifted feedback cavity (FSFC) to circulatelight through a gain medium. Light circulating through the FSFC isfrequency shifted by an acousto-optic modulator (AOM) upon each passthrough the cavity. A unique characteristic of this cavity is that,unlike a Fabry-Perot cavity, it does not selectively attenuate signalfrequencies. In the thesis “A New Method for Generating Short OpticalPulses,” applicant describes how an optical signal propagating through aFSFC is spread in frequency to generate broadband lasing, where theamount of frequency spreading is proportional to the number of timesthat light circulates through the cavity. In the Applied Physics Lettersarticle “Optical Pulse Generation with a Frequency Shifted FeedbackLaser,” applicant describes an interference condition in which thebroadband output of the laser produces short optical pulses, which havea frequency that is related to the RF shift frequency of the AOM. Thetime-domain characteristics of these optical pulses are similar to RFpulse-radio emissions.

Although pulse-radio systems are well known in the art, they are notwell suited for commercial applications. Pulse-radio is a time-domainsystem that produces broadband radiation as a natural artifact resultingfrom the generation of short-duration pulses. Broad bandwidth, hencelarge effective processing gain, makes pulse radio ideal for covertcommunications. However, its broad bandwidth, particularly the portionoccupying the low-frequency ranges of the RF spectrum, makes proposedcommercial pulse-radio systems unlikely candidates for FCC approval. Theshort pulse width of pulse-radio signals makes Rake reception verydifficult. A Rake receiver used in a pulse-radio system would require anextraordinary number of taps, on the order of the pulse repetition ratedivided by the pulse width.

SUMMARY OF THE INVENTION

Therefore it is the principle object of the present invention to providea novel and improved method and apparatus for the generation ofexcitation signals across an antenna array to control a directional beampattern. The foregoing is accomplished by repeatedly circulating orotherwise reflecting within a cavity a continuous wave through a pieceof equipment adapted to bring about a frequency shift upon each passthrough the cavity. Consequently, each of a plurality of waves insidethe cavity is provided with an incremental value of delay and anincremental frequency shift that is proportional to the amount of delay.A plurality of transmit signals are coupled out of the cavity and arewavelength demultiplexed to produce a plurality of separated transmitsignals representing different incremental delays. Each of the separatedsignals is coupled to an array element and down-converted fortransmission to produce the array's beam pattern. Similarly, an incidentRF receive signal is received by each antenna array element andconverted into one of a plurality of optical receive signals having anincremental frequency associated with the array element. The opticalreceive signals are coupled into the cavity and thus delayed andfrequency shifted in the manner previously described. A portion of thedelayed and frequency-shifted optical receive signals are coupled out ofthe cavity and separated by frequency in a receiver. Each frequencyrepresents an incremental time sample of the signal received from adirection determined by the value of the incremental delay provided bythe cavity.

It is, therefore, a second objective to provide a Rake-type receiverthat is capable of sampling signals in time without the use of delaytaps.

Another objective of the invention herein disclosed and claimed is toproduce a train of ultra-short RF pulses from a rheterodynedfrequency-shifted feedback cavity, thus providing a novelspread-spectrum communications format:

Accordingly, another object is to provide pulsed output resulting frominterference between a plurality of carrier waves, the carrier wavesbeing chosen with respect to frequency band constraints.

Still another objective is to provide a spread-spectrum format that iseasily adapted to adaptive array processing in a multiple-accesscommunications system.

Another object of the invention is to control the direction of anantenna beam pattern of an array comprising a large number of antennaelements by controlling the incremental delay of excitation of the arrayelements through the use of a single delay device.

Still another objective is that of providing a broadband antenna arraypattern having lowered sidelobe magnitudes and no secondary main-lobestructures.

An additional objective is that of shaping time-domain pulses byapplying a tapered window function to carrier-signal amplitudes in thefrequency domain.

Further objects of the invention herein disclosed and claimed are toprovide narrow pulse widths and lower time-domain sidelobes by utilizingnon-redundant spacing of interfering carrier frequencies, to shape andcontrol fringes in time-domain pulses via frequency domain adjustmentsof carrier signals, and to smooth out comb structures in the frequencydomain by dithering the frequency of each carrier signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of an array processor including a traveling-wavecavity that contains a frequency-shifting device through which opticalsignals are circulated.

FIG. 2 is a plot of time-domain pulses resulting from constructiveinterference between 10 incrementally spaced-in-frequency carriersignals.

FIG. 3 is a plot of time-domain pulses resulting from constructiveinterference between 50 incrementally spaced-in-frequency carriersignals.

FIG. 4 is a comparison plot of a time-domain pulse and a signalresulting from a sum of carrier signals whose phases have been adjustedby a maximal direct sequence code.

FIG. 5 is a plot of a time-domain signal resulting from a sum of carriersignals whose phases have been adjusted by a maximal direct sequencecode within a narrow time interval centered around the amplitude of thepulse shown in FIG. 4.

FIG. 6 is a frequency versus amplitude plot of a tapered distribution ofcarrier signals.

FIG. 7A is a plot of a time-domain pulse train resulting from the sum of19 incrementally spaced-in-frequency carrier signals having uniformamplitude.

FIG. 7B is a plot of a time-domain pulse train resulting from the sum of19 incrementally spaced-in-frequency carrier signals that are tapered inamplitude.

FIG. 8 is a plot of five of ten incrementally spaced-in-frequencycarrier signals and a signal representing the sum of the ten carriersignals.

FIG. 9A is a frequency versus amplitude plot of a group of five carriersignals having discrete incrementally-spaced frequencies.

FIG. 9B is a plot of a time-domain pulse train created by themode-locked sum of the carrier signals shown in FIG. 9A.

FIG. 10A is a frequency versus amplitude plot of two groups of carriersignals having discreet incrementally-spaced frequencies, the groupsbeing separated from each other in the frequency domain.

FIG. 10B is a plot of a time-domain pulse train created by themode-locked sum of the carrier signals shown in FIG. 10A wherein thepulses contain fringes.

FIG. 11A is a frequency versus amplitude plot of two groups of carriersignals having discreet incrementally-spaced frequencies, the groupsbeing separated from each other in the frequency domain by an amountthat is greater than the group separation shown in FIG. 10A.

FIG. 11B is a plot of a time-domain pulse train created by themode-locked sum of the carrier signals shown in FIG. 11A wherein thepulses contain fringes.

FIG. 12A is a frequency versus amplitude plot of three groups of carriersignals having discreet incrementally-spaced frequencies, the groupsbeing separated from each other in the frequency domain by an amountthat is equal to the group separation shown in FIG. 10A.

FIG. 12B is a plot of a time-domain pulse train created by themode-locked sum of the carrier signals shown in FIG. 12A wherein thepulses contain fringes and sidelobes.

FIG. 13 is a time-domain plot illustrating periodic pulses resultingfrom the sum of a plurality of incrementally spaced discrete-frequencycarrier signals.

FIG. 14 is a time-domain plot resulting from a phase-locked sum of aplurality of non-incrementally spaced-in-frequency carrier signals.

FIG. 15 is a time-domain plot showing sidelobes surrounding a pulseresulting from the sum of a plurality of incrementally spaceddiscrete-frequency carrier signals.

FIG. 16 is a time-domain plot showing amplitude-reduction in thesidelobes surrounding a pulse resulting from a phase-locked sum of aplurality of non-incrementally spaced-in-frequency carrier signals.

FIG. 17 is a plot of an antenna beam pattern for each of tenincrementally spaced discrete-frequency carrier signals.

FIG. 18 is a plot of an antenna beam pattern resulting from the sum often incrementally spaced discrete-frequency carrier signals.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

The standard method for sustaining laser oscillation uses feedback froma Fabry-Perot cavity. The multiple reflections within the cavity lead todestructive interference for all frequencies of light except thosediscreet frequencies that correspond to the standing waves of thecavity. This is demonstrated by frequency discrimination that occurswithin an etalon. The intensity of light that is transmitted through anetalon is sharply peaked at the resonance of the cavity. Non-resonantwaves destructively interfere within the cavity, thus canceling almostentirely. Therefore, a Fabry-Perot cavity used as a feedback cavity in alaser causes the laser output power to be distributed in a narrowspectral region that corresponds to the modes of the cavity. Atraveling-wave FSFC laser, as described in the cited papers co-authoredby applicant, does not selectively attenuate frequencies. Rather, thislaser is characterized by its unusually broad spectral output, which hasno mode structure. A frequency-shifting device, such as an AOM, is usedinside the cavity to incrementally shift the frequency of circulatinglight upon each pass through the cavity. A gain medium inside the cavitymaintains a constant intensity of the light over a broad spectral range,and a system of mirrors is used to circulate light through the AOM andthe gain medium. Spectral analysis of the laser output indicated acontinuous distribution of energy, which has a full-width half-maximumof 8 Angstroms centered at 5900 Angstroms.

An optical processor for an antenna array 150 shown in FIG. 1 derivesits operational characteristics from a traveling-wave FSFC 100. Theprocessor includes an injection source 110 for generating an opticaltransmit seed signal. The injection source 110 is optically coupled tothe FSFC 100. The injection source 110 may use any type oflight-emitting source to generate the transmit seed signal. In thisembodiment, the injection source includes a laser source 112 and a lasersource controller 114. The FSFC 100 includes a frequency-shifting device(such as an AOM 107) and a cavity-length adjustment device (such as atranslation stage 109), which is controlled by a scan controller 149.The FSFC 100 may also include a gain medium (not shown). Anoptical-to-RF signal converter such as a heterodyne detection device120, is optically coupled to the FSFC 100. The heterodyne detectiondevice 120 includes an output-beam wavelength demultiplexer (such as adiffraction grating 122), a fiber optic array link 124, an opticalreference source 121, a reference beam fiber optic link 123, and aphotodiode array 126 comprised of a plurality of photodiodes. Atransmit/receive coupler array 130 connects the antenna array 150 to thephotodiode array 126 and to an RF-to-optical signal converter 142 insidean optical receiver network 140. The RF-to-optical signal converter 142is coupled to the FSFC 100 via an optical beam combiner 144. The opticalreceiver network 140 also includes a receive-beam wavelengthdemultiplexer, such as receiver diffraction grating 146, coupled to theFSFC 100. The receiver diffraction grating 146 is also optically coupledto a receiver 148.

The antenna array 150 shown in FIG. 1 includes five array elements 150A,150B, 150C, 150D, and 150E. However, the present invention is capable ofcontrolling a much larger number of array elements. Broadbandcharacteristics of the FSFC 100 make it an ideal device for providingincremental delays to a very large number of elements. For example, aFSFC 100 having the same broadband characteristics as thefrequency-shifted feedback laser described in the aforementioned paperswould be capable of providing incremental delays to more than 6000antenna array elements.

The laser source 112 may be any type of laser-beam generator that canprovide beam intensities sufficient for operation of the processor asdescribed in this application and may include more than one laser. Thelaser source is preferably a semiconductor laser. The laser source 112emits an optical transmit seed signal that is coupled into the FSFC 100.For beam-forming applications, it is preferable that the transmit seedsignal be a narrow-band signal. It is possible and in some casespreferable for the laser source 112 to emit multiple optical signals,each having a different frequency. Each frequency of the transmit seedsignal emitted by the laser source 112 and coupled into the FSFC 100 isultimately used to control at least one RF beam pattern radiated by theantenna array 150. The multiple optical signals may control multiplebeam patterns and/or multiple sub-arrays of the antenna array 150.However, the embodiment of the array processor shown in FIG. 1 is usedto describe how the processor functions with respect to a singlefrequency of light input into the FSFC 100. In this case, the lasersource 112 is modulated by the laser-source controller 114 at a datarate corresponding to an information signal to be transmitted. Varioustypes of modulation may be used to produce a modulated transmit seedsignal, such as AM, FM, PAM, PSK, FH, and time-offset modulation.

The FSFC 100 shown in FIG. 1 includes a traveling-wave cavity comprisinga plurality of mirrors 101, 102, 103, and 104 for circulating opticaltraveling waves around a closed loop. Equivalently, any type ofwaveguide that utilizes optical reflection or refraction may be used tocirculate light. A travelling wave cavity may have a number ofreflecting surfaces, each of which must be properly aligned for properoperation. Thus it is preferable to use a cavity design that minimizesthe difficulty in achieving proper alignment. In this case, we assumethe FSFC 100 has a round-trip length L that is independent of thefrequency of the circulating optical waves. This causes the transmitseed signal in the FSFC 100 to be incrementally delayed relative to thenumber of round trips made by the transmit seed signal, thus providing aplurality of delayed transmit signals. The first five delayed transmitsignals are S_(Tn) (n=1, . . . , 5): S_(T1), S_(T2), S_(T3), S_(T4), andS_(T5). Each of the delayed transmit signals S_(Tn) incurs anincremental delay t_(dn)=nL/C as it circulates through the FSFC 100. Cis the speed of light in the FSFC 100 and n is an index representing thenumber of times that the transmit seed signal has circulated through theFSFC 100. The FSFC 100 is unusual in that, unlike a Fabry-Perot cavity,it does not produce modes. Thus, the length of the FSFC 100 does notlimit the frequency of the light circulating inside the cavity 100.

The FSFC 100 focuses light into the aperture of the AOM 107. Althoughthe AOM 107 shown is classified as a “bulk” optic frequency shifter,other types of frequency shifters (such as fiber optic frequencyshifters) may be used. The AOM 107 comprises a transducer 106 that iselectrically coupled to an RF source 108. Light incident on the AOM 107is split into two clusters of beams, one of which is undiffracted andone of which is diffracted and Doppler-shifted by an amountcorresponding to a shift-frequency f.sub.s equal to that of an RF signalgenerated by the RF source 108. The undiffracted and diffracted beamsemerging from the AOM 107 are spatially separate, the angular separationbetween the two beams being equal to twice the Bragg angle. In thiscase, the AOM 107 functions as an input coupler to optically couple thetransmit seed signal into the FSFC 100. However, other methods ofcoupling the transmit seed signal into the FSFC 100 may be used withoutdeparting from the scope of the invention.

The transmit seed signal coupled into the FSFC 100 from the laser source112 circulates through the FSFC 100 in a clockwise direction. As opticalsignals circulate through the FSFC 100, they are frequency shifted by afixed amount upon each pass through the AOM 107. The delayed transmitsignals S_(Tn) are frequency shifted by the shift frequency f_(s) uponeach pass through the AOM 107 that results in the signal S_(Tn) beingdiffracted. Therefore, the delayed transmit signals S_(Tn) are alsoreferred to as frequency-shifted transmit signals. The number of timeseach of the delayed transmit signals S_(Tn) has circulated through theFSFC 100, hence its delay, is implicitly known from its wavelength. Eachof the delayed transmit signals S_(Tn) has a frequency equal tof_(o)+(n−1)f_(s), where f_(o) is the frequency of the transmit seedsignal. The process of adjusting the effective length L of the FSFC 100adjusts the incremental delay without significantly affecting thefrequency or the amplitude of the optical signals circulating inside theFSFC 100. Thus, length adjustment of the FSFC 100 results in scanningthe antenna array's 150 beam pattern.

The translation stage 109 is attached to mirrors 103 and 104 of the FSFC100 and may be used to adjust the incremental delays provided by theFSFC 100 to the delayed transmit signal beams, which ultimately controlexcitation of the elements of the array 150. Other types ofcavity-length adjustment devices may be used. For example, anintracavity delay device, such as a rotating quartz block (not shown)may be used to scan the beam pattern of the antenna array 150.Frequency-selective delay may be applied to signals inside the FSFC 100or following the output of the FSFC 100 in order to providenon-incremental delay that enhances focusing capabilities of the antennaarray 150.

“A New Method for Generating Short Optical Pulses” explains that thebandwidth of the frequency-shifted feedback laser is limited due togeometric constraints of the FSFC 100. The AOM 107 diffracts light anamount that depends on the wavelength of the light causing light that isnot inside the FSFC's 100 bandwidth to transit out of the cavity 100.Either or both the placement and the selection of mirror parameters(size and focal length) for mirrors 101, 102, 103, and 104 in the FSFC100 can be used to increase the bandwidth of the FSFC 100. This alsoindicates that there are certain spatial relationships associated withthe wavelength of the circulating light within the FSFC 100 that may beused to provide variable delays to the signal beams. These spatialrelationships may also be used to provide amplitude control to thedistribution of transmit signals over the antenna array 150. Forexample, masks (not shown) and spatially selective attenuation or gaindevices (not shown) may be used to adjust the amplitude distribution ofsignals circulating inside the FSFC 100 for the purpose of beam shaping.

Losses in the FSFC 100 result in a reduction in the intensity of opticalsignals as they circulate through the cavity 100. Therefore it may beadvantageous to provide a gain medium (not shown) inside the FSFC 100.The gain medium (not shown) may be excited optically, electrically, orchemically to stimulate its initial emissions. Furthermore, the gainmedium may be the coupling means through which the transmit seed signalfrom the injection source 110 is coupled into the cavity 100. Onepossible embodiment, although not described in detail in thisdiscussion, involves the excitation of the gain medium being directlymodulated, such as by the laser source controller 114, for generating amodulated transmit signal. A first criteria of the gain medium is thatthe initial stimulated emissions are narrow band and single frequency.Accordingly, it is advantageous that the gain medium (not shown)comprises a Fabry-Perot cavity having a harmonic response that generatesmodes associated with the shift frequency f_(s) generated by the AOM107. A second criteria for the gain medium (not shown) is that itproduce narrow-band single-frequency stimulated emissions in response tonarrow-band single-frequency delayed transmit signals S_(Tn) in the FSFC100.

In the embodiment shown in FIG. 1, the AOM 107 acts as an output couplerfor coupling a portion of the delayed transmit signals S_(Tn)circulating in the cavity 100 to the heterodyne detection device 120.Other types of output couplers may be used, such as partiallytransmitting mirrors (not shown) and beam-splitters (not shown). Anoutput coupler should be chosen based on the effect that it has on theintensity of light circulating inside the cavity 100. In this case, theAOM 107 also acts as an input coupler for coupling light from the lasersource 112 into the FSFC 100. The beam from the laser source 112 isfocused on the AOM 107 at the Bragg angle so that undiffracted light iscoupled into the FSFC 100. The diffracted light represents an insertionloss. Another option involves coupling the light from the laser source112 through the AOM 107 at an angle that couples diffracted light intothe FSFC 100. Thus, the undiffracted light would represent an insertionloss. The AOM 107 may be a multichannel AOM (not shown) used for morethan one transmit or receive signal. The multi-channel AOM (not shown)may have parallel inputs that employ multiple acousto-optic modulatorson a common acousto-optic medium, and may have multiple independentchannels.

The AOM 107 and the FSFC 100 may also be used to process atransmit/receive pair of signals for full-duplex operation. This isillustrated in FIG. 1 in which a wavelength-multiplexed receive signalfrom the optical receiver network 140 is also coupled through the AOM107 into the FSFC 100. Once again, the AOM 107 functions as an inputcoupler. Other types of input couplers may be used. The choice of inputcoupler should depend on the input coupler's effect on the intensity oflight circulating in the FSFC 100 as well as the insertion lossassociated with coupling light into the FSFC 100.

The process of wavelength demultiplexing the frequency-shifted transmitsignals S_(Tn) that are output from the FSFC 100 achieves separation ofthese signals relative to their delay. S_(Tn) coupled out of the FSFC100. The coupled-out portion of frequency-shifted transmit signalsS_(Tn) are spatially demultiplexed by the diffraction grating 122, whichseparates these signals according to their wavelength into a pluralityof wavelength-demultiplexed transmission signals S_(DTn). Thus thewavelength-demultiplexed signals S_(DTn) are effectively separated withrespect to index n. Although the diffraction grating 122 is shown, othertypes of demultiplexers, such as photo-refractive elements (not shown),may be used to demultiplex signals S_(DTn).

Each of the demultiplexed transmission signals S_(DTn) is coupled intoan optical fiber that is part of the fiber optic array link 124. Eachfiber in the fiber optic array link 124 is preferably of an incrementalor uniform length to provide incremental or uniform delay to each of thedemultiplexed signals S_(DTn) and is terminated in a respectivephotodiode in the photodiode array 126.

The optical reference source 121 generates a plurality of opticalreference signals S_(Refn). Each of the signals S_(Refn) is coupled to arespective photodiode in the photodiode array 126 via the reference beamfiber optic array link 123. The optical reference signals S_(Refn) aredistinguished from each other by the index n. In this case, thereference signals S_(Refn) generated by the optical reference source 121have incremental frequencies f_(Refn)=f_(c)+nf_(s) relative to index nand have an incremental value substantially equal to the AOM's 107 shiftfrequency f_(s). A constant center frequency f_(c) is a component ofeach of the reference frequencies f_(Refn). The reference source signalsS_(Refn) are combined with the demultiplexed transmission signalsS_(DTn) with respect to the index n at the photodiode array 126. Eachphotodiode of the photodiode array 126 detects the interference betweenone of the optical reference signals S_(Refn) and one of thedemultiplexed transmission signals S_(DTn). Each of the photodiodesgenerates a corresponding radiative transmit signal S_(TXn), which inthis case is an RF transmit signal or an intermediate-frequency signal.The radiative transmit signal S_(TXn) has a differential frequencyf_(d)=|f_(c)−f_(o)|. The value of the differential frequency f_(d) maychange with respect to changes in the signal frequency f_(o). In thiscase, the reference source 121 may comprise an array of lasers (notshown) or a Fabry-Perot laser (not shown) that outputs a plurality ofmodes corresponding to the frequency shifts f_(g) generated by the AOM107.

The RF transmit signals S_(TXn) are coupled to the antenna array 150 bythe transmit/receive coupler array 130, which operates in either or botha transmit mode and a receive mode. In the transmit mode, thetransmit/receive coupler array 130 couples the RF transmit signalsS_(TXn) to the antenna array 150. In the receive mode, thetransmit/receive coupler array 130 couples RF receive signals S_(RXn)received from the antenna array 150 to the optical receiver network 140.Each of the RF transmit signals S_(TXn) is amplified by one of aplurality of amplifiers (not shown) in the antenna array 150 to generatean amplified RF transmit signal component. Each RF transmit signalcomponent is radiated by one of the plurality of array elements 150A to150E. The component radiated by each array element has an incrementaldelay defined by the index n, the round-trip length L of the FSFC 100,and the relative path length of the delayed transmit signal coupled fromthe FSFC 100 to the antenna element of the array 150. The direction ofthe radiated RF transmit signal is determined by the incremental delaysat the antenna elements.

A plurality of RF signals S_(RXn) are generated by the array elements150A to 150E, which are responsive to incident RF radiation. Each of theRF signals S_(RXn) corresponds to one of the array elements 150A to 150Eas represented by the index n. The RF signals S_(RXn) are routed throughthe transmit/receive coupler array 130 to the optical receiver network140. The RF-to-optical converter 142 converts each of the received RFsignals S_(RXn) into an optical receive signal S_(Ron). Each opticalreceive signal S_(Ron) has a unique base frequency that corresponds tothe particular antenna element that is associated with that signal. Theamplitude, phase, and frequency of the optical receive signals S_(ROn)are responsive to modulations of received RF carrier frequencies.

In the case where the antenna array 150 is a phased array (the elements150A to 150E are separated by a uniform distance), the difference infrequency between the optical receive signals S_(ROn) corresponds to aninteger multiple, such as the index n, of the AOM 107 shift frequencyf_(o) In this particular example, the frequency f.sub.n of each opticalreceive signal S_(ROn) is the sum of a base frequency f_(o) and theinteger multiple (n−1) of the shift frequency f_(s). The base frequencyof the optical receive signal S_(RO1) corresponding to the RF signalreceived by antenna element 150A has a value: f₁=f_(o). The opticalreceive signal S_(RO2) corresponding to the RF signal received byantenna element 150B has a base frequency: f₂=f_(o)+f_(s). The opticalreceive signal S_(RO3) corresponding to the RF signal received byantenna element 150C has a base frequency: f₃=f_(o)+2f_(s). The opticalreceive signal S_(R04) corresponding to the RF signal received byantenna element 150D has a base frequency: f₄=f_(o)+3f_(s). The opticalreceive signal S_(RO5) corresponding to the RF signal received byantenna element 150E has a base frequency: f₅=f_(o)+4f_(s). Each of theoptical receive signals S_(ROn) is combined in the optical beam combiner144 to produce the combined receive beam. The combined receive beam isthe sum over all n of the optical receive signals S_(Ron). The combinedreceive beam is coupled into the FSFC 100 through the AOM 107, whichdiffracts a portion of the combined receive beam into the FSFC 100. Aspreviously discussed, other types of optical couplers could be used.

The angle of incidence of the combined receive beam at the AOM 107allows a diffracted portion of the combined receive beam to be coupledinto the FSFC 100. The diffracted combined receive beam circulates theFSFC 100 in a counter-clockwise direction, and it is frequency shiftedto create a plurality of frequency-shifted receive signals S_(Rmn) as itis diffracted through the AOM 107. An index m indicates the number ofround trips in the FSFC 100 made by each signal S_(Rmn). The number offrequency shifts experienced by each signal S_(Rmn) is one less thanindex m. The index n indicates that the signals S_(Rmn) have a pluralityof discreet frequencies related to the number n of discreet frequenciesin the optical receive signal S_(Ron). An undiffracted portion of eachfrequency-shifted receive signal S_(Rmn) is coupled out of the cavity100 by the AOM 107 and separated from light emitted by the laserassembly 110 by a beam-splitter 116. Thus, a plurality of output receivesignals S′_(Rmn) having the same frequency profile as signals S_(Rmn) iscoupled into the optical receiver network 140. The receiver diffractiongrating 146 separates the output receive signals S′_(Rmn) into aplurality of component receive signals S_(Cj). Each of the componentreceive signals S_(Cj) has a different frequency represented by an indexj where j=(m−1)+(n−1). One or more of the component receive signalsS_(Cj) are coupled into the receiver 148.

The output receive signal has component frequencies f_(Cj) starting atf_(C0)=f_(o) and increasing in incremental steps j of the shiftfrequency f_(s): f_(Cj)=f_(o)+jf_(s). For example, component receivesignal S_(C4) has a frequency f_(C4)=f_(o)+4f_(s), which corresponds tosignals S′_(R51), S′_(R42), S′R_(R33), S′_(R24), and S′_(R15). SignalS′_(R15), represents the optical receive signal S_(RO1), whichcorresponds to the excitation of antenna element 150A. The signalS_(RO1) had a base frequency of f_(o) before being circulated throughthe FSFC 100 five times and consequently being frequency shifted by anamount 4f_(s). Likewise, signal S′_(R42) represents optical receivesignal S_(RO2) from antenna element 150B after it has circulated throughthe FSFC 100 four times. Signal S′_(R33) is a portion of signal S_(R03)from antenna element 150C that has circulated through the FSFC 100 threetimes. Signal S′_(R24) represents the signal S_(RO2) from antennaelement 150D that has circulated through the FSFC 100 twice. SignalS′_(R15) indicates part of the signal S_(RO1) from antenna element 150Ethat was not diffracted by the AOM 107 on its single round trip throughthe FSFC 100.

Each of the component receive signals S_(Cj) represents an incrementaldelay between each of the antenna elements 150A to 150E, where jindicates a different uniform delay that is distributed evenly acrossthe antenna array 150. This enables the receiver 148 to function as aRake receiver, thereby sampling in incrementally-spaced time intervalsby tuning to a selection of incrementally-spaced (in j) componentreceive signal frequencies f_(Cj). Each of the component receive signalsS_(Cj), when sampled simultaneously, is a sample of a specificincrementally-spaced time interval. Unlike a conventional Rake receiver,which uses a clock to time the intervals in which samples are taken, thereceiver 148 can use a plurality of frequency filters (not shown) tosample in the time domain.

The amount of incremental delay between the antenna elements 150A to150E determines the angular orientation of the array's 150 beam pattern.Adjusting the length of the FSFC 100 changes the effective viewingdirection of the antenna array 150. Thus, the process of adjusting thecavity length to scan the antenna array can be controlled by a scancontroller 149, which may be coupled to the receiver 148. In thisconfiguration, the scan controller 149 measures the receiver's 148output and uses that measurement to control the scanning process inorder to optimize the receiver's 148 reception of a particular receivedsignal. Thus, direction-of-arrival determination of received signals canbe handled efficiently and with minimum computational complexitycompared to conventional scanning techniques.

Adjusting the FSFC 100 length changes the relative delay between thecomponent receive signals S_(Cj). If the receiver 148 is operated in aRake receiver mode (more than one signal S_(Cj) being observed), thetime between samples can be adjusted by adjusting the FSFC 100 length.

The reference source 121 may include a narrow-band single-frequencyoptical signal source. This causes the radiative transmit signal S_(TXn)to be incremental in frequency with respect to index n and the shiftfrequency f_(s). This type of radiative transmit signal generatestime-domain pulses by utilizing carrier interference multiple access(CIMA), a type of spread spectrum that makes use of interference betweenmultiple carrier signals to create an information signal. Thisparticular type of CIMA is similar to mode locking in that mode-likecarrier signals having incremental frequencies are phase locked toproduce constructive interference within a given time interval,resulting in sinc-type pulses. Although the system in FIG. 1 is shown asa preferred embodiment of the invention for generating CIMA signals,other types of RF systems as well as optical systems may be used togenerate CIMA signals.

Mode locking is a technique wherein a plurality of frequency-shiftedoptical signals are summed according to a particular phase relationshipin order to produce short optical pulses. This is typically performedusing a Fabry-Perot laser whose modes are the frequency-shifted signals,and the boundary conditions of the standing wave cavity provide thephase relationship between the modes needed to generate pulses. Thepulses occur at a repetition rate equal to the shift frequency f_(s).The pulse width is inversely proportional to the number of modes N. Thepulse height (peak power) is the product of the average power of themodes and the number of modes N. In this example, there are Nequal-amplitude modes. Thus the general expression for the electricfield at a particular point in space:

${e(t)} = {\sum\limits_{{- {({N - 1})}}/2}^{{({N + 1})}/2}{E_{n}{\exp\left\lbrack {{{i\left( {\omega_{o} + {n\;\omega_{s}}} \right)}t} + {\phi(\tau)}} \right\rbrack}}}$which can be written as:

${e(t)} = {E_{o}{{\exp\left( {{\mathbb{i}\omega}_{o}t} \right)}\left\lbrack \frac{\sin\left( {N\;\omega_{s}{t/2}} \right)}{\sin\left( {\omega_{s}{t/2}} \right)} \right\rbrack}}$where E_(o) is the electric strength of each constant-amplitude mode, tis time, ω_(o) is the center frequency, and ω_(s) is the angular shiftfrequency: ω_(s)=2πf_(s)=2π/τ. τ is the period of the pulses.

The equations for the electric field strength e(t) describe amode-locked laser output. The term “mode locking,” as referred to lasersrefers to the process by which resonant longitudinal modes of a lasercavity are synchronized in phase, so as to produce a train ofelectromagnetis pulses in the laser output. However, this type of pulsedoutput can also be generated by the antenna array 150 in FIG. 1. A novelaspect of the optical processor shown in FIG. 1 is that it generates amode-lock-type spread-spectrum output comprising multiple RF carrierfrequencies that interfere to generate a baseband information signal.The sinc-type pulses produced by this invention are similar to thetime-domain output of pulse radio signals. However, the novel benefitsof the present invention are defined by the frequency domaincharacteristics of CIMA pulses. For example, a pulse-radio output thatoccupies 2 GHz of frequency spectrum occupies the spectrum from 0 to 2GHz. However, it is possible to select a group of signals in anyfrequency band to produce CIMA signals. The interference relationshipbetween the group of signals determines the time-domain characteristicsof the pulse. Thus, a CIMA output that occupies 2 GHz of bandwidth maycomprise signals in the frequency spectrum between 28 and 30 GHz.

A plot of a mode-locked output generated by the sum of tenequal-amplitude modes is shown in FIG. 2, and a plot of a mode-lockedoutput produced by the interference between 50 equal-amplitude modes isillustrated in FIG. 3. Each mode has a frequency that equals the sum ofa base frequency f_(b) and an integer multiple i (i=1, . . . , N) of anincremental separation frequency f_(i). IN the case where the FSFC 100is used to generate the modes, the base frequency f_(b) may correspondto the optical transmit seed signal's frequency, and the separationfrequency f_(i) may correspond to the shift frequency f_(s) of the AOM107. In this case, the base and separation frequencies f_(b) and f_(i)have relative values of 1000 and 0.5, respectively, and have units ofinverse time scaled by an arbitrary multiplier. The ten modes thatcomprise the pulses shown in FIG. 2 range in frequency from 1000.5 to1005. The frequency spectrum occupied by the pulses shown in FIG. 3includes 50 discrete frequencies in the range of 1000.5 to 1025. Thepulses are essentially envelopes the enclose a signal that has afrequency that is approximately the value of f_(b). The significance ofthis example is that it shows that modes can be selected from limitedfrequency spectrums to produce short time-domain pulses for CIMA.

FIG. 2 and FIG. 3 show that as the number of modes increases, the pulseheight increases and the pulse width decreases. The pulses represent aconstructive interference condition between the modes, which occurs in anarrow (t=1/τ) time domain, whereas the sidelobes indicate aquasi-orthogonal condition between the modes that exist throughout therest of the time domain. In this quasi-orthogonal region, the amplitudesand phases of the individual modes are such that they combinedestructively and thus substantially cancel.

FIG. 2 and FIG. 3 shows that as the number of modes increases, the pulseheight increases and the pulse width decreases. The pulses represent aconstructive interference condition between the modes, which occurs in anarrow (t=1/τ) time domain, whereas the sidelobes indicate aquasi-orthogonal condition between the modes that exists throughout therest of the time domain. In this quasi-orthogonal region, the amplitudesand phases of the individual modes are such that they combinedestructively and thus substantially cancel.

Because of the quasi-orthogonal nature of the modes, the application ofa pseudo-random code to control the relative phases of the modes, as isdone in OFDM-CDMA, has some inherent problems if it is applied tomode-locked signals. This is because the quasi-orthogonal nature of themodes, which is utilized to effect multiplexing, has a tendency to bedisrupted by the imposition of another coding sequence. FIG. 4illustrates a comparison between a mode-locked signal M1 and a summedOFDM-CDMA signal O1. The mode-locked signal M1 comprises the sum of 21modes, The OFDM-CDMA signal O1 comprises the sum of the same 21 modes inwhich the phase of each mode has been bi-phase shift key (BPSK) phaseshifted according to a maximal sequence of 21-chip length. As expected,the OFDM-CDMA signal O1 at time=0 is much smaller than the mode-lockedpulse of signal M1. At time=0, the mode-locked signal M1 is the sum ofthe maximum amplitudes of each mode. Because the maximum amplitude isthe same for each mode, the applied maximal sequence reduces the time=0amplitude of O1 to one, an amount corresponding to the differencebetween the number of 0 degree and 180 degree phase shifts dictated bythe maximal sequence. Thus, modes having a tapered amplitude most likelyproduce an OFDM-CDMA signal O1 having poorer pulse-reduction at time=0.More importantly, the OFDM-CDMA signal O1 exhibits spikes at other timeintervals, which add significant levels of interference to users tunedto those time intervals. These spikes represent where thequasi-orthogonality of the modes, represented by the low-amplitudeprofile of the mode-locked signal M1, has been compromised by themaximal coding sequence.

One solution for reducing the deterioration of the quasi-orthogonalnature of the modes is to select a coding sequence that is appliedwithin discreet discrete time intervals. Preferably, these timeintervals are about the size of the pulse-width of the mode-lockedsignal M1 or smaller. Because an orthogonality condition already existsbetween the modes, it is a preferred embodiment of the invention toprovide direct-sequence coding to the modes only during a discrete timeinterval that contains the pulse of the mode-locked signal. FIG. 5 showsa mode-locked signal M2 that has a 21-chip maximal code applied to itwithin a time interval that is approximately the pulse width of M2 andis centered at time=0. It is recommended that the direct-sequence codebe chosen to compensate for any amplitude tapering of the modes.

The expression for the electric field e(t) shown in FIG. 2 and FIG. 3 isa window-response function in the time domain. The electric field e(t)is simply the discrete-time Fourier transform (DTFT) of a rectangularwindow sequence w(n) in the frequency domain. The window sequence w(n)describes the amplitude distribution of the n=1 to N modes, which so farhave been considered to be equal amplitude. As N increases, the heightof the first sidelobe approaches a constant value of −13.56 dB of themain-lobe amplitude. The sidelobes contain the harmonic power of theexcitation sequence w(n), thus reducing the rise and fall rates of thesequence w(n) with respect to n reduces the sidelobe level.

The sidelobe level can be reduced by using a window sequence that taperssmoothly toward zero at the ends of the sequence. In antenna arrayprocessing, a spatial domain technique known as “array tapering” is usedto reduce sidelobes of an antenna array's beam pattern. In DS-CDMA, atechnique called minimum shift keying is used to shape chips in the timedomain in order to reduce harmonic interference in the frequency domain.Consequently, an object of this invention is to taper the amplitudeprofile of modes that have discreet frequencies in order to reducesidelobes of the mode-locked pulses in the time domain.

FIG. 6 is a plot of the spectral profile of a sequence of modes w(n)having incremental frequency spacing and amplitude tapering toward theedges of the sequence. All of the tapered window-filter techniquesreduce sidelobes at the expense of increasing the main-lobe width. Forexample, the generalized Hanning window can be interpreted as a class ofwindows obtained as a weighted sum of a rectangular window and shiftedversions of the rectangular window. The shifted versions add tighter tocancel the sidelobe structure at the expense of creating a broader mainlobe. Some other types of tapered window sequences used in finiteimpulse response (FIR) filter design that are also applicable to thepresent invention include triangular (Bartlett), Hamming, Kaiser,Chebyshev, and Gaussian windows. In the case where the excitationdistribution sequence w(n) is controlled within the FSFC 100 (forexample, this would be done in an active FSFC 100, which contains a gainmedium), a frequency-discrimination device may be used, such as a thinetalon (not shown) or an optical filter (not shown), that providesvariable attenuation with respect to wavelength. Also, a spatial filter(not shown) or mask (not shown) may be used inside the cavity 100 toattenuate certain frequencies of light relative to their spatialrelationships inside the cavity 100. The optical-to-RF signal convertermay use a window filter to taper the optical distribution input into theconverter or taper the RF distribution or the RF signal output from theconverter. Other window filters such frequency-selective or spatiallyselective variable gain or other forms of amplitude control may beapplied to signals after being coupled out of the cavity 100.

FIG. 7A illustrates the time-domain output of part of a pulse traingenerated by a rectangular frequency-versus-amplitude window of 19incrementally spaced-in-frequency modes. The pulse amplitude and timeaxes represent arbitrary units. The modes are centered at a normalizedfrequency of f_(b)=1000 and have an incremental normalized frequencyseparation of f_(i)=1. Thus, each pulse appears as an envelope functionaround a (relatively) high-frequency carrier signal. FIG. 7B illustrateshow a tapered window applied to the 19 modes reduces sidelobe height andexpands the main-lobe width of the pulses for the same distribution ofmodes shown in FIG. 7A.

FIG. 8 shows five of the ten incremental-frequency carrier signals thatcombine to create a time-domain pulse. The pulse occurs betweenarbitrary time indices of −1 and 1, centered at time=0. At time=0, eachof the carrier signals has a maximum, thus resulting in apulse-amplitude maximum. Elsewhere, such as at time=2, the difference infrequency between the carrier signals results in time offsets betweenthe maxima of the carriers. This causes the carriers to combinedestructively, resulting in an approximately null signal. Therefore, areceiver that is tuned to these carrier signals does not detect a signal(except perhaps some residual sidelobe radiation) even though thecarriers exist. Also, the receiver may be tuned to a tapered frequencyresponse around the center of the carrier frequencies in order to reducethe sidelobes of the received pulse.

The time-domain length of each carrier signal may be longer than thewidth of each time-domain pulse because the cancellation of the carriersin the time intervals between each pulse results in a substantially nullresponse from the receiver. In the system shown in FIG. 1, thetime-domain length of the frequency-shifted transmit signals S_(Tn)output from the FSFC 100 is determined by the pulse length of thetransmit seed signal. One advantage to using relatively longcarrier-signal lengths compared to the pulse width is that although thereceiver response is null, a time sampling of each of the carriersignals in the null time intervals yields amplitude and timinginformation about the pulse. A receiver may provide a predetermineddelay to each carrier signal that it receives in order to extract apulse signal from the carriers. The delay between carriers depends onthe difference in wavelength between the carriers scaled by the absolutevalue of time away from the time=0. For carriers having incrementalfrequencies f_(i) and base frequency f_(b), the difference in wavelengthbetween adjacent carriers is:λ_(i) =cf _(i)/(f _(b) +f _(i)),where c is the speed of light.

If the base frequency f_(b) is very large with respect to the bandwidth(which is defined by N·f_(i), where N is the number of carrier signals),λ_(i) is substantially constant with respect to i, and the delay isapproximately incremental. The FSFC 100 could be used to applyincremental delays to the received carrier signals. The concept of usinga predetermined delay relationship between received carrier signals thathave limited time-domain lengths is important in that it provides ameans of multiple access based on timing of the carrier signals anddelay profiles of the receivers. This technique also reduces receptionof communication signals by broadband receivers that do not apply delayrelationships to received carrier signals.

In the case where each carrier signal is transmitted by a single elementof the array 150, the delay relationships between the carriers receivedat a remote location also depends on the azimuth angle of the remotereceiver relative to the array 150. Thus, a receiver tuned to apredetermined delay relationship between the carrier signals does notreceive information embedded in the delay relationship between thecarriers if the transmitting array 150 is located at a different azimuthangle than to that which the receiver is tuned. This provides anadditional level of source verification and anti-spoofing to thecommunication-security protocol.

CIMA utilizes the TDM protocol for multiple access. In the system shownin FIG. 1, the transmit seed signal has its delay set to provide apredetermined phase relationship between the transmitted carrier signalsin one or more specific time slots. CIMA has an advantage over DS-CDMAin that the TDM aspect of CIMA limits co-channel interference toneighboring time slots, whereas DS-CDMA users distribute co-channelinterference over the entire channel. Accordingly, a CIMA receiver (suchas receiver 148) may receive its intended signal at an assigned timeslot and sample the signals in adjacent time slots to produce acancellation signal, which cancels signals that leak from the adjacenttime slots. Because interference between the carriers results in thebaseband information signal, the receiver 148 design is much simplerthan a conventional receiver design because it does not require ademodulation system.

In addition to TDM, the channel capacity of CIMA may be expanded by theuse of frequency-division multiplexing. Because the baseband informationsignals result from interference between carrier signals having aspecific mode structure and phase relationship, any group of carriersignals that exhibit the same mode structure and phase relationship maybe used to carry the information signals. This means that although twoor more information signals may occupy the same time slot, they may beseparated by the frequencies of their corresponding carriers. Thus areceiver equipped with frequency-selective filters accepts only thoseCIMA signals that are intended for that receiver. Furthermore, thesignal levels at the outputs of the filters may be monitored andselectively attenuated to reduce narrow-band interference or may beselectively enhanced to compensate for absorptive losses in thetransmission path.

Time-offset multiplexing (TOM) may be used as a form of TDM. In TOM,baseband information is correlated to a predetermined constellation oftime offsets that occur around a given time interval. The time-domainlocation of received pulses is mapped to the constellation of offsets todecipher the data sent. One advantage of TOM is that pulses that aretime offset tend to help smooth out the comb-like structure in thefrequency spectrum that results from periodic pulses in the time domain.

CIMA transmissions are controlled in the frequency domain. Because CIMAdepends on the relationship of carrier-signal phases and frequencieswith respect to each other, it is possible to smooth out thefrequency-spectrum comb structure by pseudo-randomly dithering thefrequency of each of the carrier signals. In the system shown in FIG. 1,each of the frequency-shifted transmit signals S_(Tn) coupled out of theFSFC 100 may be similarly dithered in frequency by a multi-channel AOM(not shown) that applies a pseudo-random frequency shift with respect totime to all of the frequency-shifted transmit signals S_(Tn).

FIG. 9A shows the relative frequency spectrum of a group of fivediscrete-frequency modes. The modes are separated in frequency by theamount f_(i)=0.5 scaled by a constant n. The base frequency for eachmode is f_(b)=1000 scaled by the constant n. In this case, themagnitudes of the five modes are tapered in order to reduce sidelobelevels in the time-domain sum of the modes shown in FIG. 9B. The pulsesshown in FIG. 9B are envelopes that encompass a periodic signalstructure whose frequency is approximately f_(b)=1000 scaled by theconstant n. The time axis shown in FIG. 9B has an undefined scale withrespect to time due to the undefined constant multiplier n. However, theperiod of the pulses is determined by the frequency separation f_(i)between the modes. Increasing the separation between the modes resultsin decreasing the pulse repetition rate. Increasing mode separation inthe system shown in FIG. 1 is accomplished by increasing the shiftfrequency f_(s) of the AOM 107. Increasing the width of the mode groupby adding more modes decreases the width of the time-domain pulse. Thismay be accomplished by expanding the spectral capabilities of the gainmedium in the case of an active FSFC 100 or by increasing the intensityor number of transmit seed signal coupled into the FSFC 100.

FIG. 10A shows the frequency spectrum of a pair of mode groups, eachhaving similar mode separations f_(i) and base frequencies f_(b) as themodes shown in FIG. 9A. The mode groups are separated by a relativemode-group separation frequency f_(gs)=3 (n.times.Hertz). FIG. 10Billustrates approximately five fringes occurring in each of the pulseenvelopes. The fringe patterns in the pulse envelopes result frommultiple mode groups having a distinct group separation. The systemshown in FIG. 1 may be used to generate fringes within the pulseenvelopes by inserting multiple frequency-separated transmit seedsignals into the FSFC 100. This generates multiple mode groups providedthat the losses within the FSFC are great enough to significantlyattenuate the amplitudes of frequency-shifted seed signals so that thefrequency-shifted seed signals do not completely fill the spectrumbetween the frequency-separated transmit seed signals.

FIG. 11A and FIG. 11B show how increasing the group separation frequencyf_(gs) between mode groups causes more fringes to occur in the pulseenvelopes. In the system shown in FIG. 1, increasing group separationfrequency f_(gs) is accomplished by increasing the frequency separationof the transmit seed signals. It is important to note that the shape ofthe pulse envelope does not change with increasing group separation,only the number of fringes within the envelope changes.

FIG. 12A and FIG. 12B shown relative to FIG. 10A and FIG. 10B illustratethe result of increasing the number of mode groups while maintaining thegroup separation frequency f_(gs) between the groups. As the number ofmode groups is increased, the fringes within the pulse envelopes becomenarrower and fringe sidelobes appear. Tapering the amplitude of thegroups on both edges of the frequency distribution reduces the fringesidelobes at the expense of making the fringes wider.

The method of adjusting fringes that occur within the pulse envelopesprovides a means for conveying multiple data bits within each pulse.Information streams are created by controlling the frequencydistribution of the modes to convey different data bits in the output.In the cases shown in FIG. 10B through FIG. 12B, the fringe pattern issymmetrical. Summing out-of-phase pulses or fringes with the pulseenvelope to cancel predetermined fringes within the pulse envelope maybe performed to create asymmetrical fringe patterns and thus increasethe amount of data sent.

Another type of CIMA includes non-incrementally spacing the frequencyintervals f_(i) of the transmitted carrier signals. Non-incrementalspacing of the separation frequencies f_(i) includes non-redundantspacing, random spacing, and continuous frequency over selectedfrequency bands. FIG. 13 shows periodic time-domain pulses resultingfrom the sum of incrementally spaced carrier signals. The carrier signalfrequencies are phased so their maxima coincide at a given timeinterval, time=0, constructively adding to create a pulse. Theincremental spacing in frequency between the carriers causes the maximaof the carriers to coincide periodically. FIG. 14 shows hownon-incremental spacing of the carrier frequencies attenuates theamplitude of pulses that occur away from the time interval, time=0. FIG.15 shows sidelobes that occur due to minor constructive additions of theincrementally spaced carrier frequencies, whereas FIG. 16 shows areduction in the magnitude of minor constructive additions that occurwhen the carrier frequencies are not incrementally spaced. In the systemshown in FIG. 1, non-incremental spacing of the frequency-shiftedtransmit signals S_(Tn) may be accomplished by non-redundantly orrandomly spacing the transmit seed signal or by inserting a broadbandtransmit seed signal into the FSFC 100. Sending multiple or repeatedpulses requires repeating the phase relationship between the carriersignals, which results in constructive interference.

The broadband aspects of CIMA provide benefits to antenna arrays, suchas reducing much of the sidelobe magnitude of the antenna-array patternscompared to narrow-band operation and reducing secondary main lobes.FIG. 17 shows the antenna-array array patterns for ten incrementallyspaced-in-frequency carrier signals. FIG. 18 shows the antenna-arraypatterns for the sum of the ten patterns shown in FIG. 17.

In the preferred embodiment shown in FIG. 1, the FSFC 100 provides themeans for generating frequency-shifted transmit signals S_(Tn), whichare down-converted into CIMA signals and transmitted from the antenna150. This preferred embodiment demonstrates only one of many methods forgenerating CIMA signals in order to provide a basic understanding of thecharacteristics of CIMA. With respect to this understanding, manyaspects of this invention may vary, such as in accordance with themethods used to create CIMA signals. In this regard, it should beunderstood that such variations will fall within the scope of thepresent invention, its essence lying more fundamentally with the designrealizations and discoveries achieved than merely the particular designsdeveloped.

The foregoing discussion and the claims which follow describe thepreferred embodiments of the present invention. Particularly withrespect to the claims, it should be understood that changes may be madewithout departing from the essence of the invention. In this regard, itis intended that such changes would still fall within the scope of thepresent invention. It is simply not practical to describe and claim allpossible revisions to the present invention which may be accomplished.To the extent such revisions utilize the essence of the presentinvention, each naturally falls within the breadth of protectionencompassed by this patent. This is particularly true for the presentinvention since its basic concepts and understandings are fundamental innature and can be broadly applied.

1. An apparatus comprising: a multicarrier-signal generator to receivean information-modulated electromagnetic signal and to generate anelectromagnetic information-modulated multicarrier signal therefrom, theelectromagnetic information-modulated multicarrier signal having aplurality of electromagnetic information-modulated frequency modes, anda carrier-frequency selector configured to isolate at least one of theelectromagnetic information-modulated frequency modes.
 2. A methodcomprising: accepting at least one modulation signal, and generating amodulated multicarrier signal having a plurality of frequencies, saidgenerating comprising: modulating the at least one modulation signalonto a multicarrier signal to obtain a modulated intermediate signal,and filtering the modulated intermediate signal to obtain at least oneof the plurality of frequencies within at least one predeterminedfrequency band.
 3. A method comprising: generating at least one periodicsignal having a plurality of incrementally spaced-in-frequency componentsignals, at least one spaced-in-frequency component signal being withina desired frequency band, the at least one periodic signal having anamplitude that is a function of an input signal; and coupling the atleast one periodic signal to a communication channel.
 4. A methodcomprising: generating a plurality of electromagnetic carrier signalshaving a plurality of frequencies; and combining the electromagneticcarrier signals to generate a composite signal; wherein the compositesignal has an envelope signal modulating a composite signal carriersignal having a composite carrier frequency that is a function of theplurality of frequencies.
 5. The method of claim 4, wherein each of saidelectromagnetic carrier signals has an amplitude that is a function ofan information signal.
 6. The method of claim 4, wherein the envelopesignal is a function of an information signal.
 7. The method of claim 4,wherein the plurality of frequencies are within at least one predefinedfrequency window, and wherein the electromagnetic carrier signals aremodulated by a baseband information signal.
 8. The method of claim 4,wherein said combining comprises: phase-locking the electromagneticcarrier signals to produce orthogonal interference, wherein theelectromagnetic carrier signals constructively add to create aninterference information signal having a predetermined pulse widthoccurring at one or more predetermined time intervals.
 9. The method ofclaim 8, wherein said phase-locking is controlled by a basebandinformation signal to modulate the electromagnetic carrier signals. 10.The method of claim 8, further comprising: modulating theelectromagnetic carrier signals by a baseband information signal usingpulse-amplitude modulation applied to the plurality of electromagneticcarrier signals, wherein the pulse-amplitude modulation has a pulsewidth that is longer than the pulse width of the interferenceinformation signal.
 11. The method of claim 8, further comprising:modulating the electromagnetic carrier signals with respect to apredetermined pseudorandom code signal using phase-shift keying of theelectromagnetic carrier signals within a phase-shift key time intervalequal to the pulse width of the interference information signal.
 12. Themethod of claim 8, wherein said generating the plurality ofelectromagnetic carrier signals comprises: providing for a plurality ofgroups of discrete carrier frequencies, each group having apredetermined number of discrete carrier frequencies, a spacing betweenthe discrete carrier frequencies in each of the groups determining arepetition rate of the interference information signal, the number ofdiscrete carrier frequencies in the groups determining the pulse widthof interference information signals, each interference informationsignal comprising a number of sub-pulses determined by spacing betweengroups of discrete carrier frequencies, and wherein the sub-pulses havesub-pulse widths determined by a number of groups of discrete carrierfrequencies.
 13. The method of claim 8, wherein said generating theplurality of electromagnetic carrier signals comprises: tapering afrequency-versus-amplitude window of a carrier signal amplitudedistribution to reduce time-domain sidelobe energy of the interferenceinformation signal.
 14. The method of claim 4, wherein theelectromagnetic carrier signals comprise one or more broadbandcontinuous-frequency signals.
 15. The method of claim 4, wherein theelectromagnetic carrier signals have discrete frequencies.
 16. Themethod of claim 15, wherein the discrete frequencies are incrementallyseparated in frequency.
 17. The method of claim 4, wherein theelectromagnetic carrier signals have predetermined frequencies to allowfor frequency separation of the electromagnetic carrier signals withrespect to other carrier signals.
 18. The method of claim 4, whereinsaid generating the plurality of electromagnetic carrier signalscomprises: frequency-dithering the electromagnetic carrier signals,wherein an amount of frequency dithering is substantially uniform infrequency variation with respect to time.
 19. The method of claim 4,wherein said generating the plurality of electromagnetic carrier signalscomprises: time-offsetting the electromagnetic carrier signals.
 20. Themethod of claim 4, wherein the electromagnetic carrier signals areoptical carrier signals, the method further comprising converting theoptical carrier signals to RF carrier signals.
 21. A carrierinterferometry multiple access (CIMA) transmitter, comprising: a signalgenerator to generate a plurality of electromagnetic carrier signalshaving a plurality of frequencies; and a combiner to combine theelectromagnetic carrier signals to generate a combined signal thatincludes an envelope signal modulating a combined carrier signal havinga combined carrier frequency that is a function of the plurality offrequencies.
 22. The CIMA transmitter of claim 21, wherein each of saidelectromagnetic carrier signals has an amplitude that is a function ofan information signal.
 23. The CIMA transmitter of claim 21, wherein theenvelope signal is a function of an information signal.
 24. The CIMAtransmitter of claim 21, further comprising: a transmission unit totransmit combined carrier signal over a communication channel.
 25. TheCIMA transmitter of claim 24, wherein the transmission unit comprises:an antenna array.